Saint-Perersburg State Techincal Univ., Russia, e-mail guskov@radio.stu.neva.ru

Low cost mm-wave electronically scanned antennas are developed for various civilian and military applications. The antenna is based on a planar integrated ferrite traveling wave structure and is controlled by the magnetizing of ferrite elements. There are the mathematical model of antenna and the software for calculations; experimental antenna samples were tested at frequency bands 35-37 and 75 GHz, as well as 11-12 GHz. The antenna has low profile design with thickness about half-wavelength. A beam control is very simple: 1D-scanning is provided by 1 control current only, for 2D-scanning 2 control currents are enough. Depending on antenna aperture the beamwidth can be from 5⁰ to 1⁰, insertion loss 2-4 dB. The various antenna configurations are investigated. In a simple case the scanning sector is ± 20⁰, a side lobe level –12⁰ dB. It is possible to expand the scanning sector to ± 40⁰ or to reduce side lobe level below –20 dB. Also an active version of the antenna is considered, which has more high performances.

Millimeter waves (MMW) attract attention because of wide frequency band of signals and small dimensions of components. In particular, MMW antennas can have relatively non large dimensions, keeping sharp directivity of radiation. It brings together MMW and optical waves, but unlike from last ones MMW are less affected by dust, mist, snow.

This paper describes a new class of antennas – integrated phased arrays (IPA) with ferrite control [1-3], which can be used in entire MMW band. Main advantages of these antennas is low profile integrated design, very simple beam control and low cost.

Fig.1 and 2 show the design of linear and planar antenna respectively. The main antenna component is a three-layer waveguiding ferrite-dielectric-ferrite (FDF) structure. The bottom surface of the FDF-structure is metallized, on the top surface radiating dipoles are disposed. Dielectric layer unlike from ferrite layers is not solid, but it is fabricated of one (Fig.1) or several (Fig.2) parallel rods. Wires of control winding are placed in the gaps between rods. Rows of radiating dipoles are disposed strictly above dielectric rods.

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Fig.1. Linear IPA design

Fig.2 Planar IPA design

Each dielectric rod together with close-fitting areas of ferrite layers forms a waveguide. The wave propagating along this waveguide excites currents in dipoles which, in its turn, radiate a space wave. Phase shift between currents in neighbor dipoles , and also direction of maximum radiation depend on phase velocity v of the waveguide mode:

, (1)

where q – the angle between beam direction and normal (Fig.1,2) in XOZ plane (H-plane), q=c/v – moderation factor of waveguide mode (c – velocity of light), n – an integer, l – a wavelength, d_x – a distance between dipoles along x axis. Usually is about 4, so n=2.

A current in the control winding magnetizes ferrite layers in the opposite directions parallel y axis. Variations of magnetic induction in the ferrite layer changes the value q and leads to scanning of the beam in H-plane, in accordance with (1). To scan in transversal plane (E-plane) in the case of planar antenna (Fig.2) it is necessary to control phases of waves, excited at FDF-waveguide inputs. It can be made with a help of diverse means, which will be discussed further.

Not dielectric but ferrite plane rods (shorts) are disposed between ferrite layers at the FDF-structure edges, which provide closing of magnetic flux. This provides smoothing of magnetization in the top and bottom layer and along y axis, and also decreasing of current in control winding.

Due to high permeability of dielectric (e _s=37-40) the field of the waveguide mode is concentrated in the dielectric rod and near it. Thus a coupling between neighbor waveguides is negligibly small in planar antenna. Range of variation of q value caused by magnetizing of ferrite is maximum at optimum choice of dielectric thickness. It is seen from plot in Fig.3, that for l =8 mm , accordingly the scanning sector is

Fig.3. Dependence of main wave moderation factor q vs magnetic induction in ferrite layers

Total thickness of the FDF-structure is about l /5.

In conventional traveling wave antennas reflections from radiators are summing in phase, when beam is directed normally; it causes strong mismatching of the antenna, pattern diagram distortion and gain drop. It can be easily avoided in offered antennas, because ferrite has nonreciprocity property, so a wave of reverse direction propagates with other velocity (dash line in Fig.3). Relationship q(B) for both waves is about linear, with

.

So it is enough to choose such relationship between q(0) and d_x/l , that beam would not directed normally at demagnetized state (B=0). Then reflections will be summed in different phases, so their level would stay rather small.

Currents amplitudes in dipoles diminish from the antenna input, because a power of exciting wave is sequentially radiating in the space and absorbing in waveguide itself. If as usually dipole lengths are equal, then a current in n-th dipole approximately is

where a _w – own attenuation per length unit in waveguide, a _r – attenuation due to dipole radiation. Value a _r can be controlled by variation of cross dimensions of FDF-structure and also of length of dipole. There is optimum ratio , which provides minimum gain loss. This optimum ratio is about for antennas with narrow beam (1.5-2 deg.) and correspondingly large dimensions. In this case antenna efficiency is about -3 dB. Value >1 for smaller L_x, that causes efficiency raise.

Simplest version is linear or planar antenna with 1D-scanning. Feeding of linear antenna is provided by standard rectangular waveguide; planar antenna is fed by E-horn. In both cases a dielectric matching transformer is mounted at the input of each dielectric rod of FDF-structure; the transformer is made of dielectric with e =3. The worked out transformer provides VSWR less than 1.5 in a wide frequency band. To minimize reflections an absorbing material is put on dielectric rod at the opposite edge of waveguiding structure.

2D-scanning in a planar antenna can be realized by several ways. The simplest way is feeding by E-horn and pass-by phase shifters (Fig.6). These phase shifters are fabricated at the same FDF-structure, but each phase shifter is controlled separately. Length of the phase shifter is 15-20 mm. Controlling magnetic flux is closed by П-shaped ferrite elements, on which windings are reeled up. If antenna contains N rows of radiators, then N control currents are necessary to scan in E-plane and one current more – to scan in H-plane, i.e. N+1 currents are needed.

Fig.6. 2D-scanning antenna:

a)antenna design, b)phase shifter cross-section

To decrease dimensions feeding the horn can be substituted by empty waveguide with coupling holed at side wall.

The next version of antenna (Fig.7) contains electrically controlled power divider [2]. It is the same FDF-waveguide as antenna has, but instead of radiating dipoles the slot or dipole coupling elements with waveguides of antenna array are disposed on its side wall. The magnetizing current in winding of this dividing waveguide changes the phases of waves exciting the waveguides of antenna array and provides beam scanning in E-plane. Thus , 2D-scanning is realized by only two control currents. But extreme simplicity of beam control is achieved at the expense of 2-3 dB losses increasing in comparison with antenna in Fig.6.

Fig.7. Configuration of 2D-scanning 2C-controled antenna

Note, that the same methods of performance improving can be used, which were described above: redoubling of scanning sector by switching waveguide ports, diminishing of SLL and losses decreasing (or additional beam narrowing) by means of the antenna feeding from the center.

More high performance may be achieved by using active antenna version [3]. In Fig.8 the schematic diagram of such antenna and a possible practical realization are presented. The antenna consists of electrically controlled FDF-power dividing unit with slot coupling elements, MMIC amplifiers and radiating patches. Coupling slots are etched in metallization on the upper ferrite plate.

Fig.8 Active antenna: a)schematic diagram b) design

For this antenna a loss in FDF-structure becomes not essential due to amplifiers, if the amplifier gain factor is about 20 dB (or more). For example , in this case a receiving antenna has G/T ration (gain/temperature) close to D/T_a, where D is directivity ratio, T_a – amplifier noise temperature, otherwise such an antenna is equivalent to lossless one, having the same radiation pattern diagram.

Slotted connection between FDF-waveguide and microstrip line has an ability to be varied in a wide range of coupling factor by means of parameters changing. Due to this we can choose the coupling factor of FDF-waveguide with each channel such a way, that allows to realize decreasing to the edges amplitude distribution and low side lobes level. As a result combination of high antenna performance and simple beam control as well as low profile design is attained. Planar 1D-scanning is optimum by cost, because number of amplifiers is equal to number of radiator rows, i.e. is relatively small. Such antenna is convenient to use in short cm-waves.

Electrodynamic analysis includes three groups of problems.

1. Investigation of eigen-waves propagating along FDF-waveguide: determination of phase velocity and attenuation constant of every waveguide mode, and also configuration of fields with regard to magnetization of ferrite layers. To solve this problem the method of equivalent lines and Galerkin’s method were used. Then generalized reciprocity theorem was applied for determination of coupling coefficients between complex amplitudes of propagating modes and current in dipole.
1. Study of a plane wave diffraction on FDF-structure has been carried out by means of surface impedance matrix method and space harmonic decomposition method. From here the field of radiation of single dipole on the structure, its own radiation resistance and mutual impedance matrix of dipole array have been found by means of generalized reciprocity theorem.
1. Solving of mentioned above problems allows to describe processes in the antenna with a help of equivalent UHF circuit. Currents in dipoles are found by solving linear algebraic system of equations, which takes into account magnetization of ferrite layers, multimode operation of FDF-waveguide and mutual coupling of radiating dipoles. After this pattern diagram is calculated, angle position of beam and its width, gain, efficiency and other parameters are determined for given magnetization of layers

Problem of a control magnetic circuit analysis is solved separately. The aim of the solution is calculation of magnetization distribution in ferrite layers as a function of control current and residual magnetization. Here equations of magnetic circuits and method of integral equations have been used.

At present time the software package, which realizes complete mathematical model of the antenna and which provides the ability of optimum choice of the antenna parameters, has been worked out.

Typical dependence of beam position vs. control current is represented in Fig.9. Its shape is alike hysteresis curve of ferrite magnetizing. To eliminate ambiguity of beam angle position it is necessary to execute reset in saturation state (negative, fore example) before new beam set, and then to change current monotonically to given value. Then we obtain single-valued characteristic, which is shown by solid line (dash lines show beam movement at reset).

Fig.9. Beam angle position vs control current

Minimum time of switching is 5m S. During quick change-over the energy is mainly consumed by ferrite; its value does not depend on time of switching and it is equal

,

where V – volume of ferrite, H_c – coercive force, D B – total changing of magnetic inductance, which is equal on the average 2B_m, where B_m is induction of “technical” saturation. E_mean is about 2 mJ for 8 mm-wave antenna with 550 dipoles; it is 4m J per one radiating element. If beam is switched with frequency 1 KHz, than absorbed in ferrite power is 2 W; for frequency 20KHz it will raise up to 20 W.

Static beam position is maintained by flowing constant DC in control winding. The power needed for this in mentioned antenna is 2-4 W.

If frequency of switching is less than 1 KHz, then it is profitable to use regime of magnetic memory. In this case a set impulse of voltage is applied to the winding after the reset. If amplitude of this impulse is constant, then magnetic induction increment is in straight proportion to impulse duration. After the impulse end the antenna beam moves slightly back (in accordance with one of the dash lines in Fig.9 to point of intersection with ordinate axis) and stays here arbitrarily long. Control circuit does not consume power in this state.

To take into account temperature influence on beam position (Fig.9) the antenna is to be equipped with temperature sensor, which signal is put in control processor.

The described way of control can be used, when it is needed to switch beam from one given position to another arbitrary position. Regime of periodic sweeping can be realized simpler: a voltage of rectangular shape is applied to the winding; this leads to beam sweeping with almost constant speed. In this case both reset and temperature sensor are unnecessary. It is enough to provide constancy of amplitude and frequency of control voltage with amplitude of beam sweeping is being stabilized automatically.

Experimental sample of linear antenna array of 8 mm-wave band has 250 mm length and contains 63 radiating dipoles. Beam width is 2 deg., scanning sector is ± 20 deg. (H-plane), range of control current – ± 0.7 A, power consumption – about 1 W. Pattern diagrams for different control current values are represented in Fig.10. The antenna gain is 27 dB (in combination with cylindrical-parabolic reflector, which narrows down pattern diagram in E-plane to 3 deg.).

Fig.10. Measured pattern diagrams of 8 mm-wave linear array (H-plane)

A sample of 2D-scnning planar antenna of 8 mm wave band having 22´ 25=550 radiators is shown in Fig.11. The antenna dimensions are 120´ 120 mm, thickness is 6 mm (without carrying framework). Beam width is 4 deg. (H-plane)´ 5 deg. (E-plane). The antenna has two inputs, which switching allows to extent scanning sector in H-plane. There are an empty waveguide distributor with coupling holes and 22 pass-by phase shifters at every antenna input.. Width of scanning sector in each plane is 80 deg.

Fig.11. 2D-scanning 8 mm-wave planar antenna (photo), containing 550 radiators

Experimental pattern diagram of the 8 mm-wave antenna with feeding from the center (in accordance with Fig.5) is shown in Fig.12. Side lobes level is -20 dB.

Fig.12. Measured pattern diagram of 8 mm-wave antenna with central excitation

Samples of linear antennas of 4 mm and 25 mm wave bands have been also tested. The measured parameters are in accordance with calculated ones.

The following mechanical processing have been used for fabrication of laboratory samples: ferrite slabs and dielectric rods are ground with tolerance 1-3m m by thickness, then the radiating dipoles and bottom screening layers of aluminum are sprayed on preliminary polished surface. After that all elements are glued on carrying ceramic basis with thickness 2-4 mm, which has longitudinal ditches for control winding wires. Wires are reeled up before gluing upper ferrite layer.

Cost of 8 mm-wave antenna having 2 deg beamwidth can be brought to $300 ($5 per one radiating element). Cost of the planar antenna is approximately the same per one radiating element.

In future at serial fabrication it should be taken into account, that the presented class of antennas is well adapted to modern technology, such as PCB (Printed Circuit Block), MMIC (Microwave/MM-wave Monolithic Integrated Circuits), SPS (Substrate Plasma Spraying, etc. Their usage will allow to improve the antenna parameters and to decrease cost.

The presented antennas solve the problem of electrical scanning in frequency range from 10 to 90 GHz. They have low profile design, simple beam control and nowadays are most cheap among all known scanning antennas. Antennas can be realized in different versions depending on imposed requirements. Possible applications of the antenna are

• small-size radar (automobiles, ships, helicopters);
• mobile communication systems;
• altimeters and landing systems;
• systems of transport traffic control;
• systems of satellite communications.

1. Zaitsev E.F., Yavon Yu.P. et al. “MM-wave Integrated Phased Arrays with Ferrite Control”. IEEE Transaction on Antennas and Propagation, v.42, №3б, March, 1994, p..1262-1368.
1. Zaitsev E.F et. al. “MM-wave Integrated Phased Arrays with One Current and Two Current Control”, Book of Proc. Of Microwaves 1994 Conf., Oct.25-27, London, UK.
1. A.Cherepanov, A.Guskov, et. al., “Innovative integrated ferrite phased array technologies for EHF radar and communication applications, IEEE International Symposium on Phased Array Systems and Technology, Oct. 15-18, 1996, Boston, US, p..74-77.

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